An electronic filter apparatus

ABSTRACT

An electronic apparatus is described. The apparatus includes a circuit element configured to output a signal comprising a modulated frequency component. The apparatus also includes a filter arrangement comprising first, second and third notch filter arrangements, wherein each of the first and second notch filter arrangements comprise a first series inductor, and a series shunt configuration comprising a second inductor and a capacitor coupled in series and the third notch filter arrangement comprises a series inductor and a shunt capacitor, wherein each of the notch filter arrangements are configured to generate a notch in a frequency response to attenuate the output signal at a frequency of the modulated frequency component.

The invention relates to an electronic filter apparatus. In particular,the invention relates to an electronic apparatus comprising a circuitelement, such as a switching amplifier, configured to output a signalcomprising a modulated frequency component and cascaded notch filterarrangements. The signal has phase information that is sensitive toalteration, such as audio.

BACKGROUND

FIG. 1 illustrates an example of an audio pulse width modulation (PWM)amplifier 2. An audio PWM amplifier is a switching amplifier with aswitching frequency usually in the region of 20 to 40 times higher thanthe highest audio frequency of the audio signal. The amplifier includesa power stage 4. The power stage 4 includes a gate driver or switchingcontroller 6 which generates a signal comprising square pulses of fixedamplitude with a varying width and separation. It will be appreciatedthat the gate driver 6 generates a signal and an inverted and timedelayed version of the same, the protective time delay is commonly knownas dead-time. The amplifier comprises block 14, which includes anintegrator, comparator and switching frequency clock generator, with theclock typically in the form of a triangular or saw-tooth waveform. Thelow-frequency portion of the signal generated by block 14 is the audiosignal to be amplified and the high-frequency portion from the clocksource which serves to create a PWM digital signal (i.e. a signal whichswitches between two distinct, predefined voltages) when combined withthe low frequency audio signal.

The power stage 4 includes two switching devices 8, 10. The output ofthe gate driver 6 is coupled to the two switching devices in a push-pullor totem-pole arrangement. In the circuit illustrated in FIG. 1, theswitches are field effect transistors (FET) 8, 10. In the figure, theupper FET 8 is driven by the signal output by the driver 6 and the lowerFET 10 is driven by the inverted and time delayed version of the same.The two switches 8, 10 are arranged so as to be either fully on or fullyoff such that the output of the switching devices is at either +Vcc or−Vcc (i.e. the bus voltages). This type of arrangement utilises theproperty that if the push-pull output devices are either fully on orfully off they dissipate their minimum power leading to high amplifierefficiency. The output of the power stage 4 is fed back to summation 12,whereby the input audio signal is summed with the output of the powerstage 4. The summed signal is fed to block 14 to assist in compensatingfor variations in the output voltage with respect to the input voltage,due to finite and varying load configurations, and variations in busvoltage.

The output of the power stage 4 is filtered using a passive low passfilter 16 to filter out the switching frequency to allow a loudspeakerload 18 of the amplifier 2 to only see the audio signal. The standardand commonly used filter for the passive filter 16 is a two-poleinductor-capacitor (LC) filter. The commonly used component values forthe passive filter 16 are a series inductor 20 of 20 μH and a capacitor22 of 470 nF shunted to ground. The loudspeaker 18 may naturally filterout residual switching frequency energy conducted from the amplifieroutput with the audio signal. This residual energy may cause extra heatin the loudspeaker drive units, which may result in reduced linearity ofthe loudspeaker operation. This is undesirable.

The passive filter 16 provides an approximate voltage rejection of 35 dB( 1/56) at the switching frequency of the power amplifier output and mayalso be detrimental to the audio signal integrity causes 20 degrees ofphase shift at 20 kHz with an 8 ohm load, 36 degrees of phase shift at20 kHz with a 4 ohm load and 55 degrees of phase shift at 20 kHz into a2 ohm load. Loudspeaker impedance with respect to frequency curves arenot flat and the impedance may vary from 16 ohms at loudspeakerresonance/crossover to 3 ohms at DC due to the resistance of the voicecoil. These standard component values cause audio material dependant vsloudspeaker dependant phase shifts in the audio which is noticeable tothe discerning listener. Thus the quality of the audio and the listeningexperience is degraded.

Reducing the series inductor 20 to 5 μH may reduce the phase shiftsmentioned above by a factor of 4 with an 8 ohm load. The reductionbrings the loudspeaker phase-shift in line with the expectations of ananalogue amplifier of around 5 degrees at 20 kHz with an 8 ohm load. Thesound quality from the amplifier is also brought in line with theexpectations of an analogue amplifier. It is noted that the seriesinductor should not be typically reduced below 5 μH as this componentlimits/controls the peak current seen by the switching devices 8, 10.However, reducing the series inductor also reduces the roll off achievedat the switching frequency of the amplifier, such that additionalfiltering is required to achieve a suitable level of attenuation of theswitching frequency.

In this regard, it is widely accepted within the field of audio thathigh definition audio of 16 bits or higher demands a noise floor that ismore than 90 dB below the full-scale signal, to be below the perceptionof the discerning listener. Therefore, it would be desirable to providea circuit which meets the expectations of phase-shift and sound qualityof an analogue amplifier using a PWM amplifier, for example, whileachieving a suitable level of attenuation of the switching frequency.

BRIEF DESCRIPTION OF THE DRAWINGS

The present disclosure can be understood with reference to thedescription of the embodiments set out below, in conjunction with theappended drawings in which:

FIG. 1, described above, illustrates an example of an audio pulse widthmodulation (PWM) amplifier;

FIG. 2 illustrates an audio circuit;

FIG. 3 illustrates a frequency response of a passive filter illustratedin FIG. 2;

FIG. 4 illustrates a passive filter;

FIG. 5 illustrates a frequency response of the passive filterillustrated in FIG. 4;

FIG. 6 illustrates an active filter arrangement;

FIG. 7 illustrates a further active filter arrangement;

FIG. 8 is a motor controller which can be used with the filterarrangements described in association with FIGS. 2 and 4;

FIG. 9 is a system block diagram which can be used with the filterarrangements described in association with FIGS. 6 and 7;

FIG. 10 is a system block diagram illustrating an electromagnetictransmission carrier signal (e.g. RF) recovery application; and

FIG. 11 is a system block diagram illustrating an electromagnetictransmission carrier signal (e.g. optical) recovery application.

DESCRIPTION

According to a first embodiment of the invention there is provided anelectronic apparatus comprising a circuit element configured to output asignal comprising a modulated frequency component; and a filterarrangement comprising first, second and third notch filter arrangements(i.e. three cascaded filter arrangements), wherein each of the first andsecond notch filter arrangements comprise a first series inductor, and aseries shunt configuration comprising a second inductor and a capacitorcoupled in series and the third notch filter arrangement comprises aseries inductor and a shunt capacitor, wherein each of the notch filterarrangements are configured to generate a notch in a frequency responseto attenuate the output signal at a frequency of the modulated frequencycomponent.

The first series inductor of each of the second and third notch filterarrangements may be a common mode choke.

The first series inductor of the second notch filter arrangement may becoupled to the first series inductor of the first notch filterarrangement and the series inductor of the third notch filterarrangement may be coupled to the capacitor of the first notch filterarrangement.

The apparatus may comprise a fourth notch filter arrangement comprisingan inductor and a capacitor coupled in series, wherein the inductor andthe capacitor of the fourth notch filter arrangement is coupled to thefirst series inductor of the second notch filter arrangement.

The apparatus may comprise a voltage follower arranged at the output ofthe filter arrangement. The apparatus may also comprise an operationalamplifier gain stage.

The output signal may be a pulse width modulated signal, and may be anaudio signal or a motor control signal.

The modulated frequency component may be a triangular waveform.

The circuit element may be a switching amplifier, and the switchingamplifier may be a pulse width modulation amplifier.

The circuit element may be a modulated carrier signal receiver, and themodulated frequency component is a carrier signal.

Audio is a field that specialises in high integrity signal processingand reconstruction, due to the non-linear and high dynamic range andresponse of the human ear and its ability to detect phase shift andtiming differences in sounds and music, especially when an audio signalpasses through a filter or a circuit that has an altering effect on thesignal phase information. Therefore, the apparatus described herein maybe used for any application that requires high integrity signalreconstruction or recovery from a modulated signal that is sensitive tophase preservation. The phrase modulated frequency component is usedherein to describe a signal component which has been modulated whenbeing combined with an information signal (e.g. an audio signal) in anamplifier (e.g. in an audio pulse width modulation (PWM) amplifier), ora carrier wave/component which has been modulated by an informationsignal for transmission.

FIG. 2 illustrates an audio circuit 30. The audio circuit 30 is an audiopulse width modulation (PWM) amplifier.

A PWM amplifier is a switching amplifier with a switching frequencytypically in the region of 20 to 40 times higher than the highest audiofrequency of an input audio signal. The audio circuit 30 includes apower stage 4. The power stage 4 includes a gate driver or switchingcontroller 6 which generates a signal comprising square pulses of fixedamplitude with a varying width and separation. It will be appreciatedthat the gate driver generates a signal and an inverted and time delayedversion of the same, to provide output device switching protectioncommonly referred to as dead-time. The low-frequency component of thesignal generated by block 14 is the audio signal to be amplified and thehigh-frequency component of the signal is typically a triangular orsaw-tooth waveform which serves to create a digital signal (i.e. asignal which switches between two distinct, predefined voltages) whencombined with the low frequency audio signal. The high frequencycomponent is an out-of-band signal, or a signal to be modulated by theaudio signal. As is known in the art, the low frequency, audio signaland the high frequency, switching signal (i.e. out-of-band signal) arecombined using a comparator circuit.

The power stage 4 includes two switching devices 8, 10. The output ofthe gate driver 6 is coupled to the two switching devices in a push-pullor totem-pole arrangement. In the circuit illustrated in FIG. 2, theswitches are field effect transistors (FET) 8, 10. In the figure, theupper FET 8 is driven by the signal output by the driver 6 and the lowerFET 10 is driven by the inverted and time delayed version of the same.The two switches 8, 10 are arranged so as to be either fully on or fullyoff such that the output of the switching devices is at either +Vcc or−Vcc (i.e. the bus voltages). This type of arrangement utilises theproperty that if the push-pull output devices are either fully on orfully off they dissipate their minimum power leading to high amplifierefficiency. The output of the power stage 4 is fed back to summation 12,whereby the input audio signal is summed with the output of the powerstage 4. The summed signal is fed to block 14 to assist in compensatingfor variations in the output voltage with respect to the input voltage,due to finite and varying load configurations, and variations in busvoltage. Block 14 includes an error amplifier or integrator, the outputof which is coupled to a comparator which combines the audio signaloutput with a triangular or saw-tooth waveform generated by a signalgenerator, and the output of the comparator is fed to the power stage 4.

The output of the power stage 4 is filtered using a passive filter 32 tofilter out the switching frequency to allow a loudspeaker load 18 of theaudio circuit 30 to only see the audio signal.

The passive filter 32 comprises a series inductor 34 (5 μH), and a(series) shunt arrangement comprising a capacitor 38 (470 nF) and aninductor 36 (337 nH) shunted to ground. That is to say that thecapacitor 38 and the inductor 36 are coupled in series and arranged atthe signal output of the power stage 4, along with the series inductor34. A loudspeaker 18 is coupled to the series inductor 34 and groundwith the shunt arrangement coupled in parallel with the loudspeaker 18.In comparison with the passive filter 16 of FIG. 1, an inductor is addedin series with the shunt capacitor and the resonance of these twocomponents is exploited to create a notch in the filter response. Due tothe large factor between the upper frequency of the passband and thedesign selected notch frequency this is achieved without addingadditional phase shift to the audio signal.

The passive filter arrangement 32 can include the same component valuesas may be commonly used. However, it will be appreciated that these maybe changed based on the desired response. For example, a series inductor(i.e. series inductor 34) of up to 20 μH may be used. Effectively, thecomponents of a low pass filter are used in the filter arrangement, andthe low pass filter is configured to attenuate the signal, in thisexample, at a frequency that is greater than the highest typical audiofrequency (i.e. 20 kHz) and less than the switching frequency (i.e. 400kHz in this example). As described above with the series inductor 34 (5μH), in addition to the cut-off frequency of the filter arrangement 32,the filter arrangement is configured to have a low phase shift of theorder of 5 degrees, and in this example, around 5 degrees at 20 kHz withan 8 ohm load is achieved.

The shunt capacitor 38 and the shunt inductor 36 are selected to havehigh Q factor (i.e. a low internal resistance), and to satisfy theexpression F=½π√(L*C). In this expression, L is the inductance of theshunt inductor 36, C is the capacitance of the shunt capacitor 38, and Fis equal to the switching frequency of the amplifier. It is appreciatedthat the switching frequency is equal to the frequency of the saw-toothsignal, for example, and may also be referred to as the modulatedfrequency component of the signal output by the power stage 4. In thepassive filter 32 of FIG. 2, using L=336.8 nH (i.e. L=1/((2π×F)²×C)) afurther 33.6 dB rejection is achieved, totalling 61.9 dB at theswitching frequency of 400 kHz. These values have been calculated usinga loudspeaker with an 8 ohm load.

FIG. 3 illustrates the frequency response of the passive filterarrangement 32 illustrated in FIG. 2. The upper graph illustrates gainin dB and the lower graph illustrates phase shift in degrees. As can beseen from FIG. 3, there is a notch in the frequency response at 400 kHz(i.e. 20 times the upper frequency of the audio signal of 20 kHz), whichis the switching frequency (i.e. the frequency of the modulatedcomponent) of the power stage 4. Furthermore, the phase shift at 20 kHz(i.e. the upper frequency of the audio signal) is unchanged at 5 degreeswhen compared with the known passive filter 16 of FIG. 1.

It is noted that the passive filter is not derived from k of the filter(i.e. the characteristic impedance), and that the passband/terminationimpedance, the slope of the roll-off or stopband attenuation are not ofinterest by the addition of this extra pole (i.e. the notch). The onlydesign requirement is that the notch be in-line with the switchingfrequency of the power stage or the PWM modulated signal (i.e. amodulated frequency component or a carrier wave component). Themodulated frequency component may also be referred to as a modulationfrequency component.

FIG. 4 illustrates a passive filter 40. The passive filter 40 can beused as a replacement of the passive filter 32 illustrated in FIG. 2.The series inductor 34, the shunt capacitor 38 and the shunt inductor 36of the passive filter 40 are the same as those illustrated in FIG. 2.The passive filter 40 includes three further filter configurations. Inthe first additional filter configuration an additional series inductor44 a, and a series shunt arrangement comprising an inductor 52 and acapacitor 54 are provided. The first additional filter configuration iscoupled to the series inductor 34 via a resistor 42, which is added todampen the reactive components. Resistor 42 may be omitted for highpower outputs to avoid power loss. In the second additional filterconfiguration an additional series inductor 44 b, and a shuntarrangement comprising a capacitor 48 and inductor 50 are provided.Notably, the resonance effect created by capacitor 48 and inductor 50includes inductor 50 summed with inductor 44 b, so with designoptimisation of the capacitor 48 value inductor 50 may be omitted whilststill achieving the desired notch frequency Series inductors 44 a and 44b form a common mode choke (i.e. inductors 44 a and 44 b are provided bya common mode choke in which the common mode currents in the inductors44 a and 44 b flow in the same direction through each of thechoke/inductor windings). The second additional filter configuration iscoupled to the ground connection of the capacitor 38. In the thirdadditional filter configuration an additional series shunt arrangementcomprising an inductor 58 and a capacitor 60 are added. The thirdadditional filter configuration is added on the loudspeaker side of thefilter, and is coupled to the inductor 44 a of the first additionalfilter configuration and the inductor 44 b of the second additionalfilter configuration. The third additional filter configuration iscoupled to the inductor 44 a of the first additional filterconfiguration via a resistor 56, which is added to dampen the reactivecomponents. Resistor 56 may be omitted for high power outputs to avoidpower loss. A capacitor 62 is also added across the loudspeaker load tocontrol EMC at VHF.

By adding further inductors between the first filter stage (i.e. theseries inductor 34, the shunt capacitor 38 and the shunt inductor 36)and the loudspeaker (i.e. inductors 44 a and 52) and also the groundreturn (i.e. inductor 44 b), and capacitors on the loudspeaker side ofthe inductors (i.e. capacitors 48, 54, 60 and 62), further stages offiltering, for example 3, can be cascaded which causes little furtheraudio signal phase shift at 20 kHz but adds a further 3 sets ofindependently tuned resonances that can be used on an applicationspecific basis to notch out harmonics of the switching frequencyfundamental.

The capacitance values for shunt capacitors 54 and 60 should be selectedto have negligible loading of the audio signal at 20 kHz based onXc=1/(2π×F×C). Also, with a normal amplifier operational load impedanceof 8 ohms, the impedance of each capacitor at 20 kHz should be amultiple of the load impedance, in order to make this capacitor currenta fraction of the load current. If each capacitor current is 1/3 of theload current at 20 kHz then the impedance will be ˜32 ohms and themaximum limit of capacitance will be ˜220 nF (capacitors 54 and 60)based on Xc=1/(2π×F×C).

The common mode choke 44 a and 44 b is selected so that the phase shiftat 20 kHz is within desired limits for the operational load impedance.For a 1 μH series inductor (i.e. inductors 44 a and 44 b) the phaseshift is limited to 7.2 degrees at 20 kHz into the 8 ohm operationalload.

The shunt network inductors 52 and 58 are determined based on theexpression L=1/((2π×F)²×C), where F is chosen as the switching frequencyand C is the selected capacitance for the shunt leg, here as 220 nF.Thus, shunt inductors 52 and 58 for a switching frequency of 400 kHz areselected to be 720 nH.

If inductor 50 is omitted then shunt capacitor 48 creates a resonanttank with series inductor 44 b and shunt capacitor 48 value iscalculated C=1/((2π×F)²×L) , where F is chosen as the switchingfrequency and L is the inductance of series inductor 44 b. Using thesevalues and a switching frequency of 400 kHz shunt capacitor 48 would betheoretically 158.3 nF.

If inductor 50 is included and an arbitrary value of 100 nH is used, thecalculated capacitance C=1/((2π×F)²×L), where L=inductance 44b+inductance 50. Inductor 50 may be added to the circuit to providegreater control or selectivity over the frequency being notched, vs thecapacitance value for capacitor 48.

As detailed above a single LC notch filter can provide an additional 40dB attenuation to a passive LC low pass filter configuration at a designselected frequency such as 400 kHz. If a further three LC filters ofidentical values are added in parallel, the first addition gives afurther 6 dB attenuation, the second gives a further 4 dB attenuation,the third gives a further 2 dB attenuation, totalling 74 dB. The passivefilter described in association with FIG. 4 allows up to a total of fourLC notch filters (i.e. three could be used) to be used in a passivehigh-power filter network and achieve a theoretical 220 dB attenuationwith minimal in-band signal phase shift. The ability to preserve phaseis by use of a common mode choke, and its inherent property to preservesignal integrity by the action of the forward signal path field and thereturn signal path field to cancel, which minimises the inductancepresented to a forward path current that exactly matches the return pathcurrent, such as the signal. When a further three LC notch filters areadded to the load side of the common mode choke series inductances, andthe resonances are selected to combine with differing and particularportions of the common mode choke inductances, with the across line andground (commonly referred with use of common mode choke applications asX and Y) capacitances, the exceptionally high level of passive filterattenuation can be achieved by the selectively combined interaction ofthe phase lag (inductive) and phase lead (capacitive) reactances.

FIG. 5 illustrates the frequency response of the passive filterarrangement 40 illustrated in FIG. 4. The upper graph illustrates gainin dB and the lower graph illustrates phase in degrees. As can be seenfrom FIG. 5, there is a notch in the frequency response at 400 kHz (i.e.20 times the upper frequency of the audio signal of 20 kHz), which isthe switching frequency of the power stage and gives a total 220.0 dBtheoretical rejection at the switching frequency. Furthermore, the phaseshift at 20 kHz (i.e. the upper frequency of the audio signal) is 7.6degrees with an 8 ohm load.

FIG. 6 illustrates an active filter arrangement 80. The active filterarrangement 80 could be used for low power and line level filtering. Theactive filter arrangement 80 includes all the same component'sillustrated in FIG. 4 with the addition of a voltage follower 66 (i.e.an operational amplifier with its output coupled to its invertinginput). A shunt resistor 64 is provided having a value of 10k ohms, forexample, to place less loading on the filter components. An additionalseries resistor 68 is also provided having a value of 1 to 10 ohms ifrequired. The shunt resistor 64 reduces the required power rating of thefilter components but also reduces the damping of the filter.Accordingly additional series resistors can be added to control thecircuit to have the same response as is illustrated in FIG. 5.Furthermore, resistors 64 and 68 may be omitted. Indeed, resistors 42,56, 64 and 68 are selected to control the damping of high Q inductorsand capacitors, and can all be omitted if low Q parts are used.

FIG. 7 illustrates a further active filter arrangement 90. The activefilter arrangement 90 includes all the same components illustrated inFIG. 6, except the inductor 50 and the capacitor 54 are coupled to theoutput of the voltage follower 70.

The filter arrangements depicted in FIG. 6 and FIG. 7 have very closein-band responses, and the filter arrangement depicted in FIG. 6 has anin-band phase improvement of 2 degrees. However, the filter arrangementdepicted in FIG. 6 has an out-of-band response that is dependent onoperational amplifier characteristics whereas the filter arrangementdepicted in FIG. 7 response is controlled entirely by the passivecomponents. The filter arrangements depicted in FIG. 6 and FIG. 7 may befollowed by an operational amplifier gain stage to adjust the Q/width ofthe filter.

Each of the inductors mentioned herein may be implemented as a woundinductor or magnetic device, in PCB trace/track, as an electroniccircuit that produces inductance by means of phase shift control.

The examples described with respect to FIGS. 2 and 4 are pulse widthmodulation or PWM circuit arrangements. However, it will be appreciatedthat the filter arrangements described in association with FIGS. 2, 4, 6and 7 could be used in a pulse density modulation circuit or a circuitarrangement whereby a digital bit stream (e.g. direct stream digital,super audio compact disc, and line level signals) is used to switch anFET transistor, or a communication receiver circuit involving the use ofa signal (analogue or digital) modulated by a carrier wave, for example.In other words, the filter arrangements described herein may be usedwith any circuit arrangement which use a switching arrangement similarto the one illustrated in FIG. 2 or a system utilizing a modulatedcarrier wave signal.

For example, the filter arrangements described in association with FIGS.2, and 4 could be used in the circuit arrangement 202 illustrated inFIG. 8. The circuit arrangement 202 is a motor controller and includes apower stage 204, which is fed by Block 214. Block 214 includes anintegrator, comparator and switching frequency clock generator, with theclock typically in the form of a triangular or saw-tooth waveform. Thepower stage 204 includes a gate driver or switching controller 206 whichgenerates a signal comprising square pulses of fixed amplitude with avarying width and separation. It will be appreciated that the gatedriver generates a signal and an inverted and time delayed version ofthe same. The power stage 204 includes two switching devices 208, 210.The output of the gate driver 206 is coupled to the two switchingdevices in a push-pull or totem-pole arrangement. In the circuitillustrated in FIG. 8, the switches are field effect transistors (FET)208, 210. In the figure, the upper FET 208 is driven by the signaloutput by the driver 206 and the lower FET 210 is driven by the invertedand time delayed version of the same. The two switches 208, 210 arearranged so as to be either fully on or fully off such that the outputof the switching devices is at either +Vcc or −Vcc (i.e. the busvoltages). This type of arrangement utilises the property that if thepush-pull output devices are either fully on or fully off they dissipatetheir minimum power leading to high amplifier efficiency. The output ofthe power stage 204 is fed back to summation 212, whereby the inputaudio signal is summed with the output of the power stage 204. Thesummed signal is fed to block 214 to assist in compensating forvariations in the output voltage with respect to the input voltage, dueto finite and varying load configurations, and variations in busvoltage.

The output of the power stage 204 is filtered using a filter arrangement216 to filter out the switching frequency to allow a motor 218 of thecircuit 202 to only see a motor control signal. The filter arrangement216 may be one of the filter arrangements described in association withFIGS. 2 and 4 for high power conditions.

In a further example, the filter arrangements described in associationwith FIGS. 6 and 7 could be used in the circuit arrangement illustratedin FIG. 9. The circuit arrangement assists in the conversion of a lowpower PWM source, for example, into a low power analogue waveform, inorder to be processed, amplified or transduced by an analogue mechanism.In the circuit arrangement of FIG. 9 there is provided a low power PWMsource 222 (or a bit-stream or DSD source) coupled to aDigital-to-Analogue Converter (DAC) 236 that uses a fixed switchingfrequency (e.g. a Direct Stream Digital or DSD DAC) which is coupled toa filter arrangement 238 to filter out the switching frequency of thelow power PWM source 222. The filter arrangement 238 may be one of thefilter arrangements described in association with FIGS. 6 and 7. Theoutput of the filter arrangement 238 is coupled to an analogue amplifieror a digital amplifier 240, which drives the load 242. The load 242 inthis example is a loudspeaker. The active configuration illustrated inFIG. 9 may be used in low power conditions such as line level.

FIG. 10 is a system block diagram illustrating an electromagnetictransmission carrier signal (e.g. RF) recovery application. In thesystem illustrated in FIG. 10, there is provided a modulated carrierreceiver 244 as is known in the art (e.g. an antenna coupled to apre-amplifier and amplifier). The receiver 244 is coupled to a filterarrangement 246, which is tuned to remove the carrier component/signal(or the modulated frequency component) from the received signal. Thefilter arrangement 246 may be one of the filter arrangements describedin association with FIGS. 2, 4, 6 and 7. The output of the filterarrangement 246 is coupled to a post processing block 248 dependent onthe application. For example, the received signal might be a modulatedaudio signal and the post processing block is an audio amplifier drivinga loudspeaker. The system illustrated in FIG. 10 might be a digital oranalogue topology.

FIG. 11 is a system block diagram illustrating an electromagnetictransmission carrier signal (e.g. optical) recovery application. In thesystem illustrated in FIG. 11, there is provided an optical detectiondevice 252 coupled to a modulated carrier receiver 250 (e.g.transimpedance and voltage gain amplifiers) as is known in the art. Thereceiver 250 is coupled to a filter arrangement 254, which is tuned toremove the carrier wave/component (or the modulated frequency component)of the received signal. The filter arrangement 254 may be one of thefilter arrangements described in association with FIGS. 2, 4, 6 and 7.The output of the filter arrangement 254 is coupled to a post processingblock 256 dependent on the application. The system illustrated in FIG.11 might be a digital or analogue topology.

Any quoted attenuation figures are from simulation and have utilisedaccurate RF models incorporating parasitic components where applicable,therefore, carefully selected components may be close to, but may notnecessarily match, the simulated performance.

It will be appreciated that the foregoing discussion relates toparticular embodiments. However, in other embodiments, various aspectsand examples may be combined.

The following examples form part of the present disclosure:

1. An electronic apparatus comprising:

a circuit element configured to output a signal comprising a modulatedfrequency component; and

a first, series shunt configuration arranged at a signal output of thecircuit element comprising a first inductor and a first capacitorcoupled in series, wherein the first inductor and the first capacitorare configured to generate a notch in a frequency response of the first,series shunt configuration to attenuate the output signal at a frequencyof the modulated frequency component.

2. The apparatus of example 1, comprising a low-pass filter arranged ata signal output of the circuit element, wherein the low-pass filtercomprises a second, series inductor and the first capacitor of thefirst, series shunt configuration.3. The apparatus of example 1 or example 2, wherein the output signal isa pulse width modulated signal.4. The apparatus of any one of examples 1 to 3, wherein the outputsignal is an audio signal or a motor control signal.5. The apparatus of any one of examples 1 to 4, wherein the modulatedfrequency component is a triangular waveform.6. The apparatus of any one of examples 1 to 5, wherein the circuitelement is a switching amplifier.7. The apparatus of example 6, wherein the switching amplifier is apulse width modulation amplifier.8. The apparatus of example 1 or example 2, wherein the circuit elementis a modulated carrier signal receiver, and the modulated frequencycomponent is a carrier signal.9. The apparatus of any preceding example, comprising one or morefilters arranged at the signal output of the circuit element, eachfilter comprising a third series inductor, and a second, series shuntconfiguration comprising a fourth inductor and a second capacitorcoupled in series, wherein the fourth inductor and the second capacitorare configured to have a frequency response based on a frequency of themodulated frequency component.

1. An electronic apparatus comprising: a circuit element configured tooutput a signal comprising a modulated frequency component; and a filterarrangement comprising first, second and third notch filterarrangements, wherein each of the first and second notch filterarrangements comprise a first series inductor, and a series shuntconfiguration comprising a second inductor and a capacitor coupled inseries and the third notch filter arrangement comprises a seriesinductor and a shunt capacitor, wherein each of the notch filterarrangements are configured to generate a notch in a frequency responseto attenuate the output signal at a frequency of the modulated frequencycomponent.
 2. The apparatus of claim 1, wherein the first seriesinductor of each of the second and third notch filter arrangements is acommon mode choke.
 3. The apparatus of claim 1 or claim 2, wherein thefirst series inductor of the second notch filter arrangement is coupledto the first series inductor of the first notch filter arrangement andwherein the series inductor of the third notch filter arrangement iscoupled to the capacitor of the first notch filter arrangement.
 4. Theapparatus of any preceding claim, comprising a fourth notch filterarrangement comprising an inductor and a capacitor coupled in series,wherein the inductor and the capacitor of the fourth notch filterarrangement are coupled to the first series inductor of the second notchfilter arrangement.
 5. The apparatus of any preceding claim, comprisinga voltage follower arranged at the output of the filter arrangement. 6.The apparatus of any preceding claim, wherein the output signal is apulse width modulated signal.
 7. The apparatus of any preceding claim,wherein the output signal is an audio signal or a motor control signal.8. The apparatus of any preceding claim, wherein the modulated frequencycomponent is a triangular waveform.
 9. The apparatus of any precedingclaim, wherein the circuit element is a switching amplifier.
 10. Theapparatus of claim 9, wherein the switching amplifier is a pulse widthmodulation amplifier.
 11. The apparatus of any one of claims 1 to 8,wherein the circuit element is a modulated carrier signal receiver, andthe modulated frequency component is a carrier signal.